In-band on-channel digital broadcasting

ABSTRACT

A system for combining AM and FM transmissions. In-band, On-channel, FM Digital Audio Broadcast (IBOC FM-DAB) allows simultaneous transmission of DAB and FM over existing FM allocations without interfering with conventional analog FM signals. The utility of existing FM spectrum allocations is therefore enhanced.

BACKGROUND OF THE INVENTION

The present invention relates, in general, to broadcasting informationfrom a plurality of sources to one or more receivers. More particularly,the present invention is directed to broadcasting a digital informationwaveform in the same band and on the same channel with a conventionalanalog waveform.

At present, the sound quality of audio programming over commercialanalog frequency modulation (FM) broadcast facilities is significantlypoorer than that of more modern digital signal sources such as thecompact disc. A number of attempts have been made to bring the qualityof digital audio to FM broadcasting, but these attempts have usuallygiven rise to other problems which rendered them unworkable.

For example, U.S. Pat. No. 5,038,402 to Robbins discloses an apparatusand method for broadcasting digital audio over the FM broadcast band andsuggests that such digital broadcasts might be interspersed with analogbroadcasts, across the band. This patent allows the use of the FM bandfor digital broadcast, but forces the individual broadcaster to choosebetween broadcasting in digital, with better audio but a listener baseof only the new relatively scarce digital receivers, or conventionalanalog, with poorer audio quality but available to all listeners withconventional analog FM receivers. The only other alternative is for thebroadcaster to broadcast on two frequencies, one for digital and asecond for analog; however, this presents a potential problem inobtaining a license for such broadcasting from the FederalCommunications Commission (FCC). The broadcaster may or may not be ableto acquire a license to broadcast on two frequencies in a given FM radiobroadcasting market.

The FCC, in addition to licensing individual frequency bands toindividual broadcasters, is charged with allocation of frequencyspectrum to all users for all uses. One other problem presented by thesystem of the Robbins patent and others like it is that spectrum must beallocated especially for digital FM broadcast if the number of analogbroadcasters presently on the air is to remain unchanged. This problemis exacerbated by the fact that, for most of the United States and atattainable frequencies, there is no unused spectrum. In order forspectrum to be re-allocated for a new use, the FCC and the petitionersdesiring use of a frequency band must go through a protracted,uncertain, political process which will culminate in an FCC decision onhow the spectrum in question should be used.

For these reasons, there has been a need for digital audio broadcastingover the FM band which does not require a broadcaster to abandon itsinvestment in analog FM transmission equipment, require a new frequencyassignment in the existing FM band, require the listening audience todiscard existing analog FM receivers, or force the FM programbroadcasters to undergo a protracted, uncertain and expensive process toobtain a new frequency allocation.

SUMMARY OF THE INVENTION

It is, therefore, an object of the present invention to provide a systemfor digital audio broadcasting on the existing FM broadcast band. Moreparticularly, the purpose of this invention is to provide a system forIn-band, On-channel, FM Digital Audio Broadcast (IBOC FM-DAB) whichwould allow simultaneous transmission of DAB and FM over existingallocations without interfering with the conventional analog FM signals.Such a system not only would be of great value to the broadcastindustry, but the ability to multiplex supplemental message informationover conventional analog FM transmissions would be of general interestand importance from a perspective of efficient spectrum utilization. Inaddition, a system which solves the problems raised for FM DAB willinherently find applications in many other communications' environments.

Once the requirement for an in-band, on-channel digital audio broadcastis defined, a number of subsidiary problems arise in executing thatbroadcast. In particular, the digital signal and the FM program must bemodulated together in such a way that they can be demodulated and usedby various end users on the receiver end. This causes problems ofreceiver design, which are made more complex since allowance must bemade for disturbances which occur in the communications channel betweenthe transmitter and receiver. For example, in FM transmission,demodulation is often inhibited by multipath, which applies fast,undesired phase changes to FM signals thereby causing a loss of phaselock in conventional receivers which normally use "phase locked loops"(PLLs). Reacquisition of lock with a conventional PLL causes anundesired spurious time response at the demodulation output of thereceiver, which adversely affects audio quality.

In one aspect, this invention is directed to a method for introducingsupplemental programming (more messages) to a given FM broadcast byamplitude modulating the FM waveform. The supplemental amplitudemodulation is orthogonal to the initial frequency modulation so thatboth the AM and FM programs can be demodulated, either independently ortogether, without interfering with one another. This method thusprovides a vehicle for the simultaneous transmission of supplementalprogramming, such as high bit rate DAB, with existing FM over the samespectral allocation at the same time without degrading the analog FMtransmission. This invention is applicable to any system in whichadditional programming or an additional message is to be added toenhance or supplement another program which frequency modulates acarrier.

More specifically, the present invention is directed to a method andapparatus for modulating a DAB signal, consisting of 21 digitalcarriers, onto an analog FM carrier such that mirror images of the 21digital carriers are spaced in frequency on either side of the analog FMcarrier. These DAB carriers slew through the frequency band at theinstantaneous rate of the analog FM carrier. This slewing in frequencyis accomplished without causing interference between the analog FM audioand digital subcarrier signals. The analog FM program signal thenbecomes, in a sense, a carrier for the 21 digital subchannels. Byslewing through frequency, the composite signal becomes resistant tomultipath distortion. Further resistance to multipath is derived throughthe addition of a continuously transmitted wideband reference signal tothe 21 digital subchannel modulation waveforms. This reference waveformis used at the receiver as a training system for adaptive multipathequalization with quick and continuous updating. Additional resistanceto multipath distortion comes from the use of data interleaving and datacoding systems which are customarily used to detect and correct errorsin digital signal systems. As a result, the FM DAB transmission systemof the invention provides unusually good resistance to multipath induceddistortion. The FM DAB signal is modulated at a significantly lower peakpower and is positioned within the allocated spectrum mask which islicensed to each broadcaster. This allows each broadcaster to transmitthe digital signal within the licensed frequency on a single FM channel.

An important consideration in the development of an in-band, on-channelFM digital audio broadcasting (FM-DAB) system is the requirement thatthe digital signals not interfere with the analog FM signals occupyingthe same frequency allocation. An in-band, on-channel FM-DAB signalsimultaneously occupies the same frequency allocation as a conventionalanalog FM broadcast signal. The characteristics of the digital signalmust therefore be designed to prevent degradation of the analog signal.One approach to minimizing interference is to reduce the amplitude ofthe digital signal relative to the analog signal. Of course, theamplitude of the digital signal cannot be made arbitrarily small,because interference from the analog signal and thermal noise willultimately degrade the digital signal. Once the digital signal amplitudehas been reduced to the smallest possible level which maintains thedesired bit error rate over the desired coverage area, then anothertechnique must be used to ensure non-interference with the analogsignal. One such technique is to design the frequency-domaincharacteristics of the digital signal such that it is orthogonal to theanalog signal.

Digital signals occupying the same frequency allocation as analog FMsignals add a random-noise component to the received signal. A 50-dBaudio (post-detection) signal-to-noise ratio (SNR) requires between 30and 50 dB of "protection" against digital quadrature phase-shift keyed(QPSK) and quadrature amplitude modulated (QAM) signals. The widevariation in these protection ratios arises from differing test methods;a CCIR-recommended quasi-peak detection approach gives protection ratiosof 38.5 to 48.5 dB to maintain 50-dB audio SNR in the presence ofco-channel digital 256-QAM and QPSK signals. An alternative approach,using an RMS detection to measure "unweighted" SNR, yields protectionratios of 30 to 32 dB for co-channel 256-QAM digital signals, althoughsignificant audio degradation may occur at these levels.

Additional suppression of interference may be achieved by modulating thedigital signal in a way that ensures that it is orthogonal to the analogsignal. One method of achieving this orthogonality is to design thedigital signal spectrum such that it is never superimposed directly onthe analog signal spectrum. While frequency separation may not seemfeasible for in-band, on-channel systems, it may be accomplished insystems which employ frequency sliding of the digital signal. In thisapproach, the center frequencies of the digital carriers are modulatedby the FM program. This allows the correlation between the digital andthe analog signals to be minimized, which also minimizes the mutualinterference between the signals.

Of course, practical system implementations cannot be expected tomaintain perfect orthogonality, and any correction between the analogand digital signals will result in some amount of mutual interferencebetween the signals. The amount of interference will depend on theability to prevent any overlap between the analog signal spectrum andthe digital signal spectrum. By proper design and implementation of thedigital waveform, clear-channel interference suppression of 10 to 20 dBis readily accomplished. Thus the protection ratio of 40 to 50 dB may beachieved by transmitting the digital signals approximately 30 dB belowthe analog carrier level.

Minimization of the correlation between the analog and digital signalsin a clear (multipath-free) channel does not guarantee minimalinterference in the presence of multipath. If the direct-path signal ismade orthogonal to the analog signal, the delayed-path signal will notbe perfectly orthogonal, and the amount of interference between thesignals will depend on the frequency difference (which is proportionalto the delay time of the echo and FM rate). To minimize interference inthis case, the separation between the analog signal spectrum and thedigital signal spectrum must be made sufficiently large to maintainorthogonality even in high-multipath environments. For multipath delayspreads of 1 to 5 microseconds, this separation must be between 5 and 30kHz for 100 percent modulation. For extreme multipath environments, withsufficient delay spread to cause loss of orthogonality between theanalog and digital signals, the analog FM signal will be degraded by themultipath to the point at which the decrease in signal-to-noise ratiocaused by the digital signal is expected to be imperceptible at theaudio output of the receiver.

Another aspect of the invention includes provision of receivers whichwill demodulate either the analog FM program, the digital DAB program,or both. In one embodiment of the receiver, the transmitted digitalaudio signal is extracted from the standard analog FM signal on which itis carried by a programmable notch filter based upon acoustic chargedtransport (ACT) technology. The ACT-based receiver optimizes the passageof the desired (digital) signal while suppressing the undesired (analog)signal.

In addition to CD quality stereo programming and improved data services,Digital Audio Broadcast (DAB) promises to mitigate the adverse impact ofmultipath. In-band, on-channel FM DAB delivers CD-quality audio withinexisting spectral allocations, while not interfering with existing FMbroadcast reception. Most measures proposed for mitigating multipathinvolve the use of new spectrum.

Multipath is the time domain phenomenon wherein successively delayedversions of a broadcast signal arrive at the receiver simultaneously.Multipath is typically random and time variant. A multipath channel timeresponse has an associated frequency response. Multipath is usuallycharacterized in the frequency domain in terms of amplitude fade depth,spatial and temporal correlation of fade depths, and frequency coherencywhich relates to fade bandwidth. Techniques employed for mitigatingmultipath include: spread spectrum modulation; data encoding; frequencydivision multiplexing; adaptive channel equalization; and, time,frequency and spatial diversity.

Spatial diversity in the form of multiple antennas has been shown to behelpful in improving FM reception in automobiles; however, due topractical and aesthetic considerations, the use of multiple antennas hasnot been accepted by the FM radio industry and is not considered a partof the solution for multipath mitigation. Spread spectrum has beenclearly shown to alleviate multipath, but bandwidth requirements forIBOC DAB are not consistent with existing spectral allocations for FM.

Frequency diversity becomes effective against multipath as spectralseparation employed begins to exceed multipath coherence bandwidths.Urban FM multipath is thought to have coherence bandwidths in the 30 to300 kHz range, and is thought to be resistant to in-band on-channelfrequency diversity techniques; however, FCC 73.317 defines the spectralallocation for commercial FM in the United States over a 1.2 MHzbandwidth. Compliance with FCC 73.317 allows the power within 480 kHz ofthis bandwidth to reach 25 dBc. Using some of this power for IBOC DABallows for a level of frequency diversity which is exploited towards themitigation of multipath.

Adaptive channel equalization has been shown to improve multipathreception in radio systems, but the rapidly varying nature of multipathin automobiles precludes the use of conventional adaptive equalizationtechniques. Frequency division multiplexing, data encoding and timediversity complement frequency diversity measures and ACT-basedequalization techniques provide a comprehensive in-band on-channel FMDAB system with surprisingly high multipath resistance.

Three measures inherent to the modulation method are employed tomitigate multipath: a frequency slide technique, frequency divisionmultiplexing, and an ACT-based equalization technique.

"Frequency sliding" is a modulation technique in which the carrierfrequencies of a series of digital subchannels are modulated by the FMprogram. This has the effect of producing a constant frequency offsetbetween the analog-FM carrier and the IBOC digital signals. The primarymotivation for IBOC DAB frequency slide is that sliding the DAB carrierfrequencies in synchronization with the instantaneous FM signalfrequency may be used to make conventional FM detection techniquesinsensitive to IBOC DAB. The added benefit of frequency slide ismultipath mitigation. Frequency slide increases the effective IBOC DABbandwidth for multipath mitigation without increasing the IBOC DAB noisebandwidth. Frequency slide contributes a level of effective frequencydiversity against multipath.

Frequency division multiplexing is a common practice for mitigatingmultipath. The time domain advantage of frequency division multiplexingis the reduction of intersymbol interference (ISI) in each subchanneldue to multipath with respect to the ISI which would otherwise be seenby the proportionally shorter duration symbols on a single carrier. Thefrequency domain advantage of frequency division multipath is theisolation of the effects of narrowband fading to a fraction of thesubchannels. The errors induced on the affected subchannels arerecovered through data decoding.

The ACT-based equalization technique is used to compensate fornonuniform phase distortion induced by multipath across the band. Thismeasure allows for the coherent contribution of delayed signalcomponents to the digital demodulation process. All the delayed signalcomponents contribute coherently to the demodulation of each datasymbol. The processing gains are analogous to those of an ideal channelequalizer with no adaptation time.

Data encoding and error correction are the subject of substantialresearch efforts for a variety of communications and data storageapplications, and the power and efficiency of these techniques haveincreased significantly as a result of these efforts. Three measuresinherent to the data encoding technique may be used to mitigate theeffects of multipath: block coding, convolutional coding, and datainterleaving.

Block coding is used to detect and correct errors. A portion of theerrors due to narrowband fades or to temporary fades in a moving vehiclemay be detected and corrected through block coding. Block coding is alsoreferred to as "error detection and correction".

Convolutional coding provides processing gain against losses of signallevel through soft decision Viterbi decoding. Convolutional coding alsodistributes information across subchannels which adds a level ofeffective frequency diversity to the modulation. Soft decision Viterbidecoding essentially gives the decoder the demodulation information aswell as subchannel reliability information. Data is decoded according toa set of relative subchannel confidence metrics.

Data interleaving is used to distribute burst errors between levels ofcoding so as to make burst errors appear random. Although convolutionalencoding adds processing gain to the demodulation process, errors whichdo propagate through soft decision Viterbi decoding are usually burstyin nature. Interleaving spreads out burst errors in time so as to enablecorrection by the block decoder.

The DAB receiver includes a frequency tracking delay elementinterference canceler and FM demodulator which removes a dominant toneor FM interference signal by subtracting it from a delayed replica.Cancellation is maintained through a time delay that tracks theinstantaneous frequency of the dominant FM interfering waveform. A delayis generated, accurately controlled and dynamically adjusted in responseto changes in the instantaneous frequency of the dominant tone or FMinterference signal. A phase or phase threshold detector is used totrack small errors in cancellation phase in order to close the loop onthe tracking canceler and to make the tracking canceler resistant tomultipath. The control voltage at the adjustable delay element varieswith the instantaneous frequency of the predominant tone. In the casewhere an FM signal is tracked and cancelled, the control voltage becomesa demodulated FM program.

This embodiment of the receiver provides the ability to cancel or filterout a single large undesired signal whose frequency is unknown, changingor both. The tracking delay element notch filter is an adaptable filterwhich uses a very simple feed back implementation to continuously adjustthe center frequency of a notch filter in response to the instantaneousfrequency of a predominant input signal. The critical implementationcomponent is a single adjustable delay time.

Cancellation is made possible by delay elements whose relative delaysare adjusted quickly and linearly in response to the center frequencyand whose amplitude responses may also be adjusted to control the depthof the cancellation. A 180° phase detector provides the delay controlsignal, is relatively easy to implement, may be used in a thresholdsense or as a linear detector, and may be adjusted in detectionsensitivity to control the loop dynamics which establish the accuracy offrequency tracking.

The tracking notch can track the instantaneous frequency of an FM inputsignal in real time. The control voltage which causes the frequency ofthe notch to track the frequency of the predominant incoming waveformtracks the instantaneous frequency of the input signal to control theinstantaneous notch frequency. The tracking canceler demodulates FM byestimating instantaneous frequency directly with this voltage ratherthan indirectly by tracking phase as in a conventional PLL. Phasediscontinuities, such as those which may be encountered while movingthrough a multipath environment, have less effect on a tracking cancelerwhich tracks frequency than on a conventional PLL which tracks phase.The result is an FM demodulator which is more stable than a conventionalPLL in a moving multipath environment.

In another aspect of the invention, the IBOC FM-DAB receiverincorporates an FM to AM conversion canceler which functions upon theprinciple that the amplitude of the FM to AM conversion interferencecomponent is correlated to the instantaneous frequency of the FM signalas a function of the channel multipath. The FM to AM conversion cancelerestimates this correlation and continuously updates the estimate. Acancellation signal is generated within the receiver which correspondswith the correlation estimate and is used for canceling the effectivemultipath from the received signal. Three embodiments of the FM to AMconversion canceler are shown. The first provides a base band cancelsignal which is generated from a "look up table". The look up tableincludes a running estimate of the FM to AM conversion interference tobe expected from a given instantaneous FM frequency. A second embodimentbased on polynomial channel estimation includes a polynomial generatorwhich is driven by the FM signal frequency for estimation of the FM toAM conversion interference in the channel. This estimate is subtractedfrom the raw DAB composite as before. The coefficients of the polynomialare derived by measuring and integrating the cross correlation betweenthe resulting DAB composite and each term of the polynomial. In a thirdembodiment of the FM to AM conversion canceler, polynomial channelestimation is implemented in the intermediate frequency (IF) section ofthe receiver. For the third embodiment, two separate polynomials areused to continuously estimate and cancel the in-phase and quadraturecomponents of the FM to AM interference caused by multipath.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 illustrates the generation of AM over FM through amplitudemodulation of an FM waveform;

FIG. 2 illustrates the isolation of FM from AM over FM at the receiver;

FIG. 3 illustrates the use of envelope detection for demodulating the AMcomponent of AM over FM;

FIG. 4 illustrates the generation of AM over FM through frequencymodulation of an AM waveform

FIG. 5 illustrates the use of coherent detection for demodulating the AMcomponent of AM over FM;

FIG. 6 illustrates the isolation of AM from AM over FM by conversion ofAM over FM to AM over a constant frequency carrier;

FIGS. 7a, 7b and 7c illustrates the multipath induced generation ofinterfering signals from AM waveforms, FM waveforms, and AM over FM;

FIG. 8 illustrates an example of the spectrum of a signal, generated bymultipath induced FM to AM conversion, which interferes with AMdemodulation;

FIG. 9 illustrates the application of preemphasis at the transmitter tothe AM program in the interest of mitigating possible multipath inducedFM to AM conversion interference at the receiver;

FIG. 10 illustrates the application of deemphasis at the receiver to thedemodulated AM program in the interest of mitigating possible multipathinduced FM to AM conversion interference;

FIG. 11 illustrates the demodulation of both FM and AM programs by usinga tracking notch filter;

FIG. 12 illustrates the cancellation of multipath induced AM over FMcross-interference terms;

FIG. 13 illustrates a spectral representation of the plurality of DABchannels on either side of the analog FM program signal;

FIG. 14 illustrates how the DAB channels slew in frequency with the FMprogram signal;

FIG. 15 illustrates how the DAB channels slew in frequency with the FMprogram signal;

FIG. 16 illustrates the baseline IBOC FM-DAB system in block diagramform;

FIG. 17 illustrates the channelized data transmitted referencemodulator;

FIG. 18 illustrates a possible modulation symbol set for the channelizeddata transmitted reference modulator of FIG. 17;

FIGS. 19a, and 19b illustrate the spectra of data and referencesubchannels for the channelized data transmitted reference communicationsystem;

FIGS. 20a and 20b are block diagrams for a channelized data transmittedreference self-sampling receiver;

FIGS. 21a and 21b illustrate clear channel data reception through theself-sampling receiver of FIG. 20;

FIGS. 22a and 22b illustrate two path channel data reception through theself-sampling receiver of FIG. 20;

FIGS. 23a and 23b illustrate diffuse multipath channel data receptionthrough the self-sampling receiver of FIG. 20;

FIG. 24 illustrates time division multiplexing of a reference waveformwith data subchannel waveforms;

FIG. 25 is a block diagram of a channelized data transmitted referenceequalizer receiver;

FIG. 26 is a simplified delay element canceler;

FIG. 27 illustrates the time response of the canceler of FIG. 26;

FIG. 28 shows the theoretical wideband frequency response of the FIG. 26canceler;

FIG. 29 shows the theoretical narrowband frequency response of the FIG.26 canceler and gives some fundamental equations which describe canceleroperation;

FIG. 30 (a) illustrates the architecture of and FIG. 30 (b) and FIG. 30(c) illustrate two approaches for choosing tap weights for finiteimpulse response (FIR) filter variable delay elements;

FIG. 31a, FIG. 31b, and FIG. 31c gives examples of tap weight setschosen to yield different sensitivities of delay to the control signal;

FIG. 32 illustrates a tap structure where symmetry is exploited so thattap groups track in cancellation magnitude as the cancellationfrequency, through relative delay, tracks the interference frequency;

FIG. 33a and FIG. 33b illustrate circuits used in an integrated circuit(IC) canceler employing analog control of the taps of a notch filtercanceler;

FIG. 34 is a block diagram of an IC canceler having FIR based adjustabledelay groups, quadrature (≈90°) signal generation, cancellation throughlinear combination, gain adjustment for cancellation depth control,applied analog control signals, and outputs for feeding the phasedetector;

FIG. 35 is a block diagram of a phase detector designed for the trackingcanceler of FIG. 34;

FIG. 36a, FIG. 36b, and FIG. 36c illustrate timing diagrams explainingoperation of the phase detector of FIG. 35;

FIG. 37 illustrates how the outputs of the FIG. 35 phase detector may beused to generate analog control signals for the adjustable delayelements;

FIG. 38 is a block diagram for a closed loop tracking delay elementcanceler;

FIG. 39 is a block diagram of a closed loop tracking canceler employinganalog control signals for adjusting the variable delay elements;

FIG. 40 is a block diagram of a closed loop tracking canceler employingdigital control signals for adjusting the variable delay elements;

FIG. 41a, FIG. 41b, and FIG. 41c illustrate a tap weight set whichimplements a walking delay element tap group;

FIG. 42 illustrates an FM demodulator, based on variable delay elements,which is insensitive to spurious FM waveform phase variations such asdue to multipath;

FIG. 43 illustrates a cross sectional slice of an acoustic chargedtransport (ACT) tapped delay line;

FIG. 44 illustrates a transversal filter utilizing an ACT delay line;

FIG. 45 is a schematic diagram of an interference canceler;

FIG. 46 is a schematic diagram of a polynomial estimator embodiment ofan FM to AM interference canceler; and

FIG. 47 is a schematic diagram of a polynomial estimation embodiment ofan FM to AM interference canceler is used in IF processing.

FIG. 48 is a schematic diagram of an ACT-based notch filter according tothe invention.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENT

A) AM over FM composite signal generation, modulation and demodulation.

The generation of AM over FM is illustrated in FIG. 1. An FM messagem_(FM) (t) 10 excites a VCO or other FM modulator 12 to generate an FMwaveform 14. This FM waveform 14 is multiplied in mixer 16 by AM messagem_(AM) (t) 18 to yield double sideband suppressed carrier (DSB-SC) overFM signal 20. DSB-SC over FM signal 20 is summed at 22 with FM signal 14to yield double sideband large carrier (DSB-LC) AM over FM signal 24.For proper generation of DSB-LC AM over FM signal 24, the peak amplitudeof the FM 14 component should be greater than the peak value of theDSB-SC over FM component 20. The FM message m_(FM) (t) 10 is modulatedonto the frequency of the AM over FM modulation waveform 24, while theAM message m_(AM) (t) is modulated onto the amplitude of the AM over FMmodulation waveform 24. Both may be demodulated, either together orindependently, without interference from the other.

FIG. 2 illustrates a process used to isolate FM 14 from AM over FM 24. Ahard limiter 30 or zero crossing detector is used to clip the amplitudeinformation from the AM over FM waveform 24, leaving the FM waveform 14isolated for demodulation through a discriminator or phase locked loop.

FIG. 3 illustrates an envelope detector used for demodulating the AMcomponent of AM over FM 24. The AM over FM waveform is rectified bydiode 40, yielding a rectified version 42 of the AM over FM waveform. Alow pass filter 44 is used to filter out the high frequency components,leaving the AM envelope 46. The AM envelope 46 in turn is AC coupled at48 to yield the demodulated AM message 18.

An alternative method for combining FM and AM at the modulator is shownin FIG. 4. Whereas FIG. 1 illustrates the generation of AM over FM byamplitude modulating an FM waveform, FIG. 4 illustrates the generationof AM over FM by applying frequency modulation to an AM signal. Inessence, then, this embodiment involves heterodyning the AM waveformwith the FM waveform. The FM message is modulated into the FM waveform14. A fixed frequency local oscillator ("LO") signal is generated at 50,and mixed with AM message signal 18 at mixer 52 to produce a double sideband suppress carrier DAB signal 54 at the fixed LO frequency. The localoscillator signal 50 and the double side band suppress carrier DABsignal 54 are summed at 56 to produce a double side band large carrierDAB signal 57 at a stationary local oscillator frequency. The stationarylocal oscillator frequency also shifts the frequency of the FM waveform14. This is accomplished by mixing at mixer 60 the local oscillatorfrequency with the FM signal to produce intermediate signal 62. Signal62 is band pass filtered at 64 at the sum frequency to produce afrequency shifted FM waveform 66. The FM at sum frequency signal is usedto shift the double side band large carrier DAB signal 57 to thefrequency range of the original FM signal 14. This is accomplished bymixing the FM at sum signal 66 and the double side band large carrierDAB signal 57 in mixer 68, and then band pass filtering at 70 theresultant signal at the desired resulting frequency to produce a doubleside band large carrier AM over FM signal 24. The original FM signal 14can be characterized as a F₁ +δF signal. The FM at sum signal 66 canthen be characterized as F₁ +δF+F₂, where F₂ is the frequency of thestationary LO signal 50. Finally, the output signal is then F₁ +δF+F₂-F₂ which is equal to F₁ +δF. This means that the amplitude modulationis applied over the FM signal and then returned to the original FM bandof FM signal 14.

As best shown in FIG. 5, demodulation of the AM component of AM over FM24 may be realized through a coherent AM demodulator. Coherentdemodulation yields a better demodulation signal to noise ratio thanenvelope detection. For coherent demodulation, the AM over FM waveform24 is multiplied at 80 by the FM waveform 14 isolated as shown in FIG.2. The group delay in the direct AM over FM signal path going to themultiplier 80 should equal the group delay through the hard limiter 30signal path which isolates the FM waveform 14 on its way to themultiplier. Differences between these group delays yields an FMdiscrimination component in the product 82 which interferes with the AMmessage. The product 82 is passed through a low pass filter 84 whichyields AM envelope 46. AM envelope 46 is AC coupled at 86 to yield theAM demodulation waveform 18.

AM detection signal to noise ratio may be further improved by isolatingthe AM modulation component to a carrier of constant frequency. Thisprocess is shown in FIG. 6. The isolated FM signal 14 is multiplied at90 by a local oscillator (LO) signal 92 at some LO frequency f_(L). TheFM signal 14 may be thought of having instantaneous frequency f_(i) (t)defined by

    f.sub.i (t)=f.sub.c +Δf.sub.i (t)                    (1)

The product 94 is band pass filtered at 96 to isolate the sum frequencycomponent around f_(c) +f_(L). The resulting FM signal at a higher IF 98has instantaneous frequency f_(iHIGH) (t) defined by

    f.sub.iHIGH (t)=f.sub.c +f.sub.L +Δf.sub.i (t)       (2)

This FM signal at the high IF 98 is multiplied at 100 by the received AMover FM waveform 24. The product 102 is band pass at filtered 104 aroundf_(L), yielding signal 106 at the difference frequency f_(diff). Thedifference frequency f_(diff) (t) may be found by subtracting f_(i) (t),the instantaneous frequency of the AM over FM signal 24 given inequation (1), from f_(iHIGH) (t), the instantaneous frequency of the FMsignal at the high IF 98 given in equation (2). The resulting differencefrequency f_(diff) (t) is shown to be ##EQU1## The signal 106 atf_(diff) is therefore an AM signal at a carrier of constant frequencyf_(L). This AM signal may be demodulated using a conventional AMenvelope detector or a conventional AM coherent detector.

Multipath is known to induce distortion on FM signals, and manifestsitself not only in the phase of the FM signal but also in the envelopeof the FM signal. Similarly, multipath induces distortion on AM signals,which is additive to the AM signal envelope but also distorts the AMsignal phase. In the case of AM over FM, distortion terms from AM and FMboth interfere with each of the desired messages. This compoundinterference scenario is illustrated in FIG. 7. In FIG. 7a, multipath isshown to degrade FM 14 into FM plus additive interference components FM*and AM*, where * denotes interference caused by multipath originating inan FM signal. In FIG. 7b, multipath is shown to degrade AM 18 into AMplus additive interference components AM** and FM**, where ** denotesinterference caused by multipath originating in an AM signal. FIG. 7cillustrates the case of AM over FM 24, where FM degrades into FM plusFM* and AM* while AM degrades into AM plus AM** and FM**. In multipathreception of AM over FM, AM reception is degraded by interfering termsAM**, which originates in the AM signal, and AM*, which originates inthe FM signal. At the same time, FM reception is degraded by interferingterms FM*, which originates in the FM signal, and FM**, which originatesin the AM signal. The orthogonality between AM and FM signaling breaksdown in multipath. However, several measures may be taken to alleviatethis breakdown.

One method for decoupling the AM over FM interference between AM and FMis to mutually isolate the spectra of the two programs. As an example,the FM program could occupy a spectrum from DC to 60 kHz, while the AMprogram would occupy a spectrum from 75 kHz to 300 kHz. The AM and FMsignals would be isolated in their spectra over and above the isolationinherent in the orthogonal signaling (orthogonality which breaks down inmultipath).

Another method for mitigating interference due to multipath is theapplication of preemphasis and deemphasis to the AM program. This methodassumes that the AM modulation index is very small. This assumptionimplies that FM yields large multipath induced interferers FM* and AM*,while AM yields small multipath induced interferers fm** and am**. AMpreemphasis and deemphasis has the effect of mitigating AM*.

The spectrum of multipath induced FM to AM conversion interferenceresembles the spectrum 120 of FIG. 8. The FM to AM conversioninterference is more significant at lower frequencies than at higherfrequencies. FIG. 9 illustrates the application of preemphasis at thetransmitter, where a raw AM message 130 is passed through a preemphasisfilter 132 on its way to the AM modulator. The preemphasis filteramplifies the low frequencies in the message with respect to the highfrequencies in the message so as to allow the low frequencies to moreeasily overcome relatively higher levels of FM to AM conversioninterference 120. FIG. 10 illustrates the application of deemphasis atthe receiver. The raw demodulated AM message 134 is passed through thedeemphasis filter 140 which attenuates higher levels of FM to AMconversion interference 120 in the process of correcting the distortionon the message 130 induced by the preemphasis filter 132.

The application of preemphasis and deemphasis to the AM program iscompatible with the spectral isolation of the AM program from the FMprogram. Furthermore, the expected shape of the magnitude spectrum ofAM*, such as the one shown in FIG. 8, may be used to derive optimumpreemphasis and deemphasis functions, optimal cutoff frequencies forspectral isolation, or optimal combinations of both.

The AM and FM may be demodulated individually through the use of atracking notch filter as shown in FIG. 11. A tracking notch filter 150accepts AM over FM 24 as input, and tracks the instantaneous frequencyof the input 24 with the instantaneous frequency of the tracking notch150. The FM message 10 is demodulated in the process of tracking theinstantaneous frequency of the input 24. The output of the trackingnotch 150 is an AM over FM signal with the carrier removed (DSB-SC AMover FM) 20. This may be multiplied at 152 by FM only 14 to yield aproduct 154 whose baseband component, isolated through low passfiltering at 156, is the demodulated AM message 18.

Tracking notch demodulation, or signal separation in general, may beused in conjunction with cancellation techniques to mitigate theadditive interference terms induced by multipath. The AM over FM 24signal after multipath may be separated by signal separators 160 intoFM+FM*+FM** component 162 and AM+AM*+AM** component 164. In particular,AM* 163 may be removed from AM+AM*+AM** 164 and FM** 161 may be removedfrom FM+FM*+FM** 162 using cancellation techniques illustrated in FIG.12. To cancel FM** 161 from FM+FM*+FM** 162, the estimate of AM+AM** 172is used as an interference reference in an adaptive FM** canceler 166.The result, an estimate of FM+FM* 170, is applied as an interferencereference to the AM* canceler 168. The AM* canceler is an adaptiveinterference canceler designed to remove AM* 163 from AM+AM*+AM** 164.

The outputs of the cross-interference cancelers 166 and 168 are FM+FM*170 and AM+AM** 172. Conventional equalization techniques may be used toextract FM from FM+FM* 170 and AM from AM+AM** 172. Furthermore,information derived from the cross-interference cancellation processillustrated in FIG. 12 may be employed to improve the performance of theequalizers which follow it. Thus, cross-interference cancellationtechniques may restore the orthogonality between the AM and FMcomponents of the AM over FM signal.

B) IBOC DB: In-Band On-Channel Digital Broadcast

Portions of this description are written without reference to audio.However, for those portions in which audio is discussed, it is to benoted that other non-audio applications may be served by the samemethods and structures. Therefore, the acronyms IBOC FM-DAB and IBOCFM-DB are roughly interchangeable.

1) Transmission of IBOC FM-DB waveform

a) DAB waveform, designed to enable separation of the analog FM anddigital programs while both are being received in the same band at thesame time.

Turning now to a more narrow embodiment of the present invention,directed to in-band on-channel digital audio broadcast, FIG. 13illustrates digital data subchannels 220 arrayed around analog FMcarrier 210 within the spectral mask 200 licensed to a givenbroadcaster. In the context of AM over FM discussed above, the digitaldata subchannels 220 are the AM message signal 18 modulated onto the FMsignal 10. The digital data waveform frequencies are tied to theinstantaneous frequency of the analog carrier using the AM over FMmodulation method and slew with the carrier from one end of the spectralmass to another.

In FIG. 14, the digital data channels 220 have slewed across the band inlock step with the instantaneous analog FM frequency 210. In FIG. 15,the data channels 220 have slewed from end to end within the licensedspectral mask 200. The digital data channels 220 do not exceed thespectral limitations which make up part of the license.

FIG. 16 illustrates a baseline system block diagram for in-bandon-channel digital audio broadcasting in conjunction with commercial FM.FIG. 16 incorporates the concepts discussed generally as AM over FM intoa more narrowly defined in-band on-channel FM DAB embodiment.

The FM-DAB transmitter block diagram is shown in the top half of FIG.16. A DAB serial data stream 252 is demultiplexed at 254 into 21parallel subchannel data streams 256. Streams 256 are modulated intodata subchannels 260 by data modulator 258. Data demultiplexing 254 andsubchannel modulation 258 are synchronized to the 19 kHz FM stereo pilottone 250. The data modulator 258 also generates a reference signal 262which is also synchronized to the 19 kHz FM stereo pilot tone 250. Thedata modulation waveform 260 and reference waveform 262 are added at 264and their sum 266 is applied to preemphasis filter 268. The resultingDAB composite waveform 18 serves as an "AM message" for AM over FMmodulation. AM over FM is generated in FIG. 16 in the same way that itis illustrated in FIG. 1.

The FM-DAB receiver block diagram is shown in the bottom half of FIG.16. The transmitted AM over FM signal 24 is subject to multipath andadditive channel noise on its way to the receiver. The resulting inputwaveform 270 is applied to the receiver front end section 272, whichselects the appropriate channel and converts it to an IF frequency 274.The AM over FM signal 24 at the intermediate frequency (IF) 274 isprocessed by a signal separator 160, which implements tracking notchdemodulation of the FM signal 162 and the DAB (AM) signal 164, as shownin FIG. 11. The tracking notch filter in the signal separator 160 alsoapplies deemphasis, as best shown in FIG. 10, to the recovered AMcomposite. The raw demodulated baseband (BB) FM message 162 may beapplied to an FM stereo demodulator to recover the conventional analogFM audio program. Baseband FM signal 162 is also applied to clockrecovery circuit 276 to extract timing waveforms from the 19 kHz FMstereo pilot 250. The timing waveforms 278, raw demodulated FM message162, and raw digital baseband demodulated AM composite 164 are appliedto cross canceler equalizer 280.

Cross canceler equalizer 280 performs two functions. The first iscancellation of FM to AM components in the recovered AM messagewaveform. Cross cancellation is done in three steps. The first step iscorrelation of average levels of the AM message waveform 164 as afunction of instantaneous FM frequency indicated by the demodulated FMprogram 162. An averaged mapping is generated of AM interference voltageagainst FM demodulation voltage. Averaging is possible because the FMprogram is uncorrelated to the AM program. The mapping is continuouslyaveraged and updated. The second step is to estimate the FM to AMinterference signal by applying the FM demodulation voltage 162 as aninput to the mapping function. The third step is to subtract the FM toAM conversion interference estimate from the raw recovered AM signal164. FM to AM conversion interference is the result of signal multipathdistortions. FM to AM conversion interference canceling is discussed ingreater detail below.

The second function of the cross canceler equalizer 280 is multipathequalization. Multipath equalization takes advantage of the transmittedreference signal 262 included in the DAB composite 266. An adaptivelinear combiner continuously trains on the reference signal 262 which ispresent in the DAB composite 266. The equalizer implements a least meansquares (LMS) algorithm to continuously adapt its equalization operationinto synchronization with the timing waveforms derived from the 19 kHzFM stereo pilot. 250

The output of the cross canceler equalizer 280 is an equalized digitalbaseband composite signal 282. This DAB composite 282 is passed on to asubchannel matched filter bank 394 which implements matched filters forthe 21 subchannels 256. The matched filtered waveforms 284 are in turnpassed on to a bank of 21 subchannel demodulators which recover the 21parallel DAB subchannel data streams 288. These in turn may bemultiplexed into a final recovered DAB serial data stream.

The DAB serial data stream may then be processed in outboard highfidelity stereo decoding hardware for listener use.

The following broadcast parameters have been chosen for IBOC-DAB:

FM MODULATION: Corresponds to commercial FM as described in FCC 73.317.Existing FM broadcasting will not change. It should be noted thatcommercial FM broadcasts are known to far exceed the spectral maskrequirements given in FCC 73.317. This means that FM broadcasters areknown to be much more careful about their allocations than they have tobe.

AM MESSAGE: Consists of 21 subcarriers carrying a total of 399 kilobitsper second of data. The first subcarrier is at 9.5 kHz. Subsequentsubcarriers are spaced at 9.5 kHz intervals. The last subcarrier is at199.5 kHz. Added to this is a pilot waveform.

AM PREEMPHASIS: Consists firstly of weighting the power of each of the21 subcarriers by a factor of f³⁵ before modulation of the subcarriers.Consists secondly of an analog integration (take the mathematicalintegral) of the modulated composite (includes the pilot referencewaveform). The net result is that subcarrier power is weighted by f¹.65.

MODULATION SPECTRUM: Complies with FCC 73.317. The smallest DABsubcarrier, at 199.5 kHz, has -44 dBc power upon transmission. Thelargest DAB subcarrier, at 9.5 kHz, has -30 dBc power. Total AMsubcarrier power is -17 dBc. Most of this power is within 120 kHz of thecenter frequency of the allocation. FCC 73.317 requires that transmittedpower between ±120 and ±240 kHz be below -25 dBc. The scheme describedtransmits DAB power at -28 dBc, leaving a 50% margin. FCC 73.317 callsfor less than -35 dBc from ±240 kHz to ±600 kHz. The scheme describedtransmits less than -45 dBc of DAB power in this interval.

CONVENTIONAL FM DEMODULATION: Is still fully operational. DAB via AMover FM is transparent to conventional FM demodulation techniques.

C) A multipath resistant transmitted Reference Data transmission system.

In order to improve the quality and reliability of IBOC FM-DAB throughmultipath environments channelized data transmitted reference signalingmay be used. This combines data subchannelization in the interest ofincreasing the baud interval with a transmitted reference that is sharedby all data subchannels in order to equalize against multipath. StandardFM may also realize increases in multipath resistance from use of thissystem.

This system improves digital communication in the presence of multipathfor a given bandwidth efficiency and with very fast multipath adaptiontime. This system combines data subchannelization with a transmittedreference signal which is shared by all the data subchannels for fastmultipath equalization.

Channelized data transmitted reference modulation is illustrated in FIG.17. Bipolar input data 350 is applied in parallel to a number of datasubchannel waveform generators 354-358, which preferably are impulseresponse generators or finite impulse response (FIR) generators. 21subchannels are used in FIG. 17, although a greater or fewer number maybe used. Each bit of data applied to each waveform generator excites amodulation symbol specific to its respective subchannel. In every baudinterval, one modulation symbol is generated for every subchannel withits polarity determined by its respective subchannel input data. Inevery baud interval, a reference symbol (denoted "Sampler" 358 in FIG.17) is also generated with constant reference polarity. As an example, amodulation symbol set is illustrated in FIG. 18. In this example, eachof the data subchannels 360-372, which correspond to the channels at theoutput of data demultiplexer 254 of FIG. 16, is shifted slightly infrequency from the previous data subchannel. Reference subchannel 374(which corresponds to sampler 358) is wideband with respect to each datasubchannel. The reference subchannel 374 has a spectral powerdistribution that overlaps the spectral power distribution of all thedata subchannels, as best shown in FIG. 19. The reference spectrum 380of FIG. 19b spans the spectrum of all the data subchannels of FIG. 19a.

Data demodulation may be accomplished by using a self-sampling receiveras best shown in FIG. 20. In FIG. 20a, the received modulation waveform390 is applied to a bank of subchannel matched filters 394. Datasubchannels 360-372, which are narrowband with respect to the referencesubchannel 380, convolve into wide time pulses 396 as best shown in FIG.20b. The relatively wideband reference subchannel 380 convolves into arelatively narrow sampling pulse 398. The reference sampling pulse 398is multiplied at multipliers 400 by the matched filter output. Theresult 401 reflects the polarity of the subchannel data symbol whichreflects the polarity of the message data 350. A properly timedintegrate and dump operation 402 integrates the product of the matchedfiltered reference and each subchannel over time 404 (2.5 μsec) when thesampling pulse 398 is present. The polarity of the resulting integralsyields the received data in each subchannel 406.

Clear channel simulation of the integrated data output is shown in FIGS.21a and 21b. Polarity of + and - data are clearly recovered. Theadvantage of channelized data transmitted reference modulation is shownin cases where multipath is present, and an example is shown in FIG.22a. This simulation calls for a two path channel as shown in FIG. 22b.These two paths, delayed by about 10 microseconds in this example, causea channel time response which may be modeled by a pair of impulses 410.These redundant channel paths cause distortion in the matched filterresponses 396 at the demodulator. However, multiplication of thereference sampler 358 with the matched filter response correlatesmultipath delayed responses of the reference sampler with the matchedfilter respones of data subchannel matched filters. The integral of theproducts in FIG. 22b shows that this correlation allows for polarity tobe coherently restored and successful + and - detection to take place.

FIG. 23a illustrates a more extreme case of diffuse multipath 414.Although multipath is diffuse, inherent correlation of reference anddata subchannel waveforms allow for coherent detection as shown in FIG.23b at 416.

A number of variations of the reference signal transmission system orits operation are possible, among them are: mutual orthogonality betweensubchannel modulation symbols alleviates cross-talk between subchannels;data subchannel modulation symbols may consist of mutually orthogonalchirp waveforms; data subchannel modulation symbols may consist ofmutually orthogonal PN sequences; the reference waveform may consist ofa PN sequence, a chirp, a pseudorandom noise burst or any waveform thatoverlaps all the data subchannels in spectral content; and, thereference may alternatively be time division multiplexed with the dataas shown in FIG. 24.

In the preferred embodiment for channelized data transmitted referenceIBOC FM-DAB, 21 subchannel modulation waveforms are used. These arewindowed tone bursts starting at 19 kHz and spaced at 9.5 kHz. The baudinterval is 52.6 microseconds. Baud intervals are synchronized to the 19kHz stereo pilot tone. The reference is a small component of eachsubchannel tone. A block diagram of the receiver is shown in FIG. 25.Rather than using the reference in a self-sampling mode, the referencemay be used to continuously reprogram a quickly adaptive programmablemultipath equalizer 422. In addition to matched filter 424, a baudinterval circulating averager (IIR filter) 426 may be used to isolatethe reference waveform 374 for equalization training 420. Equalizationat 422 respects a timing reference 428 derived from the received 19 kHzFM stereo pilot tone 250.

Subchannelization works against multipath by extending the baudinterval. For a given data rate, the baud interval is increased inproportion to the number of subchannels chosen. Increasing the baudinterval to exceed the longest multipath delay spread is necessary forthe reference to be valid for training equalization. Equalization isdesigned to take place within a single baud interval. The reference,periodic in a baud interval, does not equalize from one baud interval tothe next. Increasing the baud interval to exceed the longest multipathdelay spread limits multipath induced intersymbol interference tosymbols in adjacent baud intervals. The number of data subchannels arechosen accordingly. For the case of in-band on-channel FM DAB, 52.6microseconds exceeds any expected multipath delay spreads. This 52.6microseconds, being the reciprocal of 19 kHz, also conveniently allowsfor synchronization to the 19 kHz FM stereo pilot 250.

Essential to multipath operation is the necessity of the reference tooccupy the same spectrum as all the data subchannels. The fact that thesame spectrum is used by the reference as the data makes the reference avalid reference waveform for sensing or characterizing the multipathseen by the subchannel data signals. In this respect, the transmittedwaveform is a phase reference waveform valid for all the datasubchannels. Channelization of the data makes possible duplicate use ofthe same reference for all data channels which improves the powerefficiency of the transmitted reference.

The use of a reference 380 which is of wider bandwidth than any of thesubchannels means that the reference correlates or match filters into anarrower pulse than any of the data subchannels. The bandwidthdifference between the reference and any data subchannel enables pulsecompression of the reference with respect to the data subchannels. Thispulse compression allows the reference to be used as a phase correctingsignal sampler for the reference channels in multipath.

D) Tracking Delay Element Notch Filter and FM Demodulator

The operation and structure of the tracking delay element notch filterand FM demodulator is now described. First, the delay element notchfilter is described. Next, the operation of the variable delay elementis treated. Then the operation of the phase detector is addressed. Thena closed loop cancellation system is discussed, including FMdemodulation.

Delay Element Notch Filter

A simplified delay element notch filter is illustrated in FIG. 26. Asingle delay element 440 provides a delay T. The signals at the input442 and at the output 444 of the delay element are added in summer 446.The signal 448 at the output of summer 446 reflects the application of aset of notches to the input signal 442.

The time impulse response of the delay element notch filter of FIG. 26is shown in FIG. 27. The impulse 450 at t=0 represents the component atthe output 448 of the summer 446 due to the input signal 442 of delayelement 440. The impulse 452 at t=T represents the component at theoutput 448 of the summer 446 due to the output signal 444 of delayelement 440. The time impulse response h_(i) (t) 454 of the delayelement notch filter 440 is

    h.sub.i (t)=δ(t)+δ(t-T)                        (4)

The frequency response H_(i) (f) of the delay element notch filter isevaluated by taking the Fourier transform of its time impulse responseh_(i) (t.sub.). The result is

    |H.sub.i (f)|=|cos (πfT)|(5)

H_(i) (f) is depicted in FIG. 28 and contains many notches, of whichnotches 460, 462, 464 are illustrated. These notches are also referredto as nulls or spectral zeroes f_(z). The first spectral zero 460 is atf_(z) =1/2/T, and the rest 462, 464, et seq. are spaced at intervals of1/T. The location of the n'th null f_(z) may be found using ##EQU2## ,where n may take on integer values of 0 and greater.

Narrowband operation of a simple delay element notch filter lends itselfto the analysis of isolated notches. The narrowband perspective of thedelay element notch response H_(i) (f) is given in FIG. 29.

Narrowband operation lends itself to simple approximations. A narrowbandapproximation for H_(i) (f) is found by taking the two term Taylorexpansion of H_(i) (f) in Eqn. 5 about f_(z) 470. The result 472, shownin FIG. 29, is

    |H.sub.i (f)|=|sin (π(f-f.sub.z)T)|≅π|f-f.sub.z |T (7)

Consider a signal of interest separated from the notch frequency f_(z)470 by a frequency offset f.sub.Δ 474, as shown in FIG. 29. Suppose alsothat the notch were positioned to within an offset of f 476 from thefrequency of the interference signal to be cancelled. Following thelinear approximation 472 of H_(i) (f), the ratio of the attenuationM.sub.Δ 478 seen by the signal of interest to the nominal attenuation M480 seen by the interference signal is equal to the ratio of thefrequency offsets 474 and 476. FIG. 29 defines relative depth 482 of thenotch, in equation form, to be ##EQU3##

Equation 8 describes the way tracking accuracy affects cancellationdepth in a tracking canceler.

FIG. 29 also illustrates limitations on tracking speed relevant to thecase of a simple digitally tracking system. In a simple digitallytracking system, the notch frequency f_(z) 470 would be changed inincrements of δf 484. The interference frequency would be measuredperiodically at some clock rate f_(cl), which would also be used toupdate f_(z) 470. If the notch frequency f_(z) 470 could be moved by theincrement δf 484 once every cycle of the clock f_(cl), then the fastestrate at which the notch frequency could be moved would be the SLEW 486,or

    SLEW=δf·f.sub.∫                        (9)

Consider that the interference frequency is detected and tracked indiscrete increments. Then f 476 in FIG. 29 would serve as a frequencyoffset threshold. If, at the detection time, the interference frequencywere offset from the notch frequency f_(z) 470 by more than f 476, thenthe notch frequency f_(z) 470 would be either incremented or decrementedby δf 484. Otherwise, if the interference frequency were offset from thenotch frequency f_(z) 470 by less than f∫₁ 476, then the notch frequencyf_(z) 470 would remain the same during the next period of the updateclock f_(cl). Under this condition, the most accurate tracking occurswhen the threshold f 476 and the increment δf 484 are related at 488 sothat

    δf=2f.sub.∫                                     (10)

Variable Delay Element

The simplified delay element notch filter illustrated in FIG. 26 employsa delay element 440 of delay T. FIG. 28 illustrates that the notchfrequency f_(z) is dependent upon the delay T. The delay T must bechanged in order to change the notch frequency f_(z). Changes Δf_(z) inthe notch frequency f_(z) are related to changes ΔT in the time delay Tthrough ##EQU4##

To change the cancellation frequency f_(z) 470, it is necessary toadjust the delay T 440. The preferred embodiment for implementing anadjustable delay T 440 between a reference point 442 and a delay point444 is to use groups of taps in a tapped delay line. One group of tapsimplements an adjustable reference group delay corresponding to 442,whereas the second tap group implements an adjustable longer overallgroup delay corresponding to 444. The relative delay T 440 between thesetwo tap groups may be adjusted by adjusting the absolute delays of eachor both delay groups with respect to one another.

A group of taps, configured as a single variable delay element, is shownin FIG. 30a. Transversal filter tap configurations which implementeasily adjustable delay elements are shown in FIGS. 30b and 30c. InFIGS. 30b and 30c, each impulse (arrow) represents the output of asingle tap in a transversal filter tap group.

A variable delay element consists of a group of closely spaced taps in adelay line 490 which together form a composite delay 492. Taps 494, 498and 504 are static, and taps 496, 500, 502, 506 and 508 are dynamic. Thestatic taps 494, 498 and 504 maintain constant tap weights, and dynamictaps 496, 500, 502, 506 and 508 change in weight, and vary the delay.

FIG. 30b illustrates a tap weight scheme for a variable delay elementemploying complementary positive dynamic taps. The three static taps 498are full amplitude. The dynamic taps 500 and 502 are always positive,and vary in a complementary fashion. "Complementary fashion" means thatif one dynamic tap has amplitude A, where 0<A<1, then the other hasamplitude 1-A. For a baseline delay, the dynamic taps 500 and 502 areset to half amplitude. For the minimum delay, the first dynamic tap 500is set to full amplitude and the second dynamic tap 502 is set to zeroamplitude. For the maximum delay, the first dynamic tap 500 is set tozero amplitude and the second dynamic tap 502 is set to full amplitude.Delay may be varied continuously by varying the dynamic taps 500 and 502continuously in this complementary fashion. As 500 is lowered, 502 israised and the delay is raised. As 500 is raised, 502 is lowered and thedelay is lowered. The delay varies very linearly with the values of thedynamic taps. Delay offset as a function of dynamic tap weight may beapproximated theoretically from the phase of the tap group with respectto the center of the tap group.

FIG. 30c illustrates a variable delay element employing dynamic taps ofopposite sign. "Opposite sign" means that if one dynamic tap implementsa weight of A, then the other dynamic tap implements a weight of -A. Thethree static taps 504 in FIG. 30c are full amplitude, and the dynamictaps 506 and 508 are always of opposite sign and may vary continuously.For a baseline delay, the dynamic taps 506 and 508 are set to zero. Forthe minimum delay, the first dynamic tap 506 is set to full positiveamplitude and the second dynamic tap 508 is set to full negativeamplitude. For the maximum delay, the first dynamic tap 506 is set tofull negative amplitude and the second dynamic tap 508 is set to fullpositive amplitude. Delay may be varied continuously by varying thedynamic taps 506 and 508 continuously in this complementary fashion. As506 is lowered, 508 is raised and the delay is raised. As 506 is raised,508 is lowered and the delay is lowered. The delay varies very linearlywith the values of the dynamic taps. Delay offset as a function ofdynamic tap weight may again be approximated theoretically from thephase of the tap group with respect to the center of the tap group.

A preferred embodiment of the tapped delay line of FIG. 30a is oneemploying an acoustic charge transport (ACT) device. ACT devices arediscussed in Hunsinger et al., U.S. Pat. No. 4,633,285, which isincorporated herein by reference.

The ACT tapped delay line may implement a programmable transversalfilter (PTF) through the addition of on-chip tap-weighting and memorycircuits and may be used as a notch filter by appropriate control of thetaps. A notch filter based upon ACT tapped delay line is shown in FIG.48. The weighting of the taps is accomplished through the use ofprogrammable attenuators that set the magnitude of the coefficient. Theaccumulation function required by the transversal filter is performed bytwo summing busses; these busses are connected to the inverting and thenoninverting inputs of an off-chip differential amplifier in order topermit bipolar tap weighting. The tap weights are stored in on-chipstatic random-access memory (SRAM). In order to change the response ofthe filter, the user simply supplies a data word containing the desiredcoefficient, the address of the tap to be loaded, and an enable signalat the time the change is desired. For further information on anACT-based PTF, see The ACT Programmable Transversal Filter, MicrowaveJournal, May 1991.

High-speed transversal filters are now available through a technologyknown as acoustic charge transport or ACT, which combines the speed andsimplicity of analog components with the programmability and delaycapability of digital processing. This technology allows the practicalimplementation of transversal filters which operate over a frequencyrange of 500 kHz to 180 MHz, providing several hundred parallel delaysover a range of several nanoseconds to several microseconds. Forapplications requiring fast updating of the filter response, ACT filterswith update times of less than 100 nsec have been demonstrated.

An ACT device is an analog delay line in which discrete samples of aninput signal are formed as a series of charge packets that propagate ina depleted transport channel induced in a GaAs substrate, as shown inFIG. 43. Hunsinger et al., U.S. Pat. No. 4,633,285 discloses anACT-based delay line and the disclosure therein is incorporated hereinby reference. These charge packets are formed and transported byelectric fields that arise from piezoelectric coupling to a propagatingsurface acoustic wave (SAW) which is generated directly on thepiezoelectric GaAs substrate. Unlike the operation of conventional SAWdevices, the ACT device employs the SAW only as a parametric pump or"clock"; all signal information is contained in the propagating chargepackets rather than in the SAW. This obviates the deleterious effects ofvarious acoustic phenomena which are observed in SAW devices, becausethese effects do not directly interact with the charge packets.

The charge packets are confined within a transport channel formed in anepitaxial surface layer having a thickness that is an appreciablefraction of the SAW wavelength at the clock frequency. This architectureisolates the charge packets from interfacial traps at the substratesurface and epitaxial layer interface. Thus, ACT devices operate in adeep "buried-channel" mode, and exhibit extremely high transportefficiencies at clock frequencies that are easily achieved in the UHFregion (300 to 1000 MHz). The excellent transport efficiency at UHFclock rates forms the basis for a variety of high speed, highperformance analog signal processors using ACT technology.

The phenomenon leading to charge packet formation in the input sectionof an ACT device results in an intrinsically high performance samplingoperation. Effective aperture times have been measured to be less thanten percent of a clock period leading to precise sample formation andthe potential for providing frequency conversion directly in the ACTdevice.

An image charge is induced in arbitrary metal features fabricated withinthe propagation path that allows the amount of charge in each packet tobe sensed without affecting the contents of the packet. This enables therealization of high performance tapped delay line structures that formthe basis for a rich set of analog signal processors.

The fabrication of ACT devices is amenable to the integration of GaAsintegrated circuit elements to provide interface and control functions.The potential for providing delay and completely integrated RF signalprocessing functions is unprecedented.

One of the most useful applications of the delay and sensing functionsinherent to the ACT device is in the transversal filter. The classicaltransversal filter is shown in FIG. 44. In this device an input signalis processed by passing it through a sequence of delay elements. Aftereach delay element, the signal is sensed, weighted by a predeterminedcoefficient, and sent to an accumulator. The original signal is passedthrough another delay element, a different weighting coefficient isapplied, and the result is sent to the accumulator. This cycle of delay,sense, weight, and accumulate is repeated in each stage of the filter.The signals from each tap add coherently when the signs of the weightingcoefficients match the phase of the input signal and a large signalappears on the output terminal of the filter.

The programmable transversal filter of FIG. 44 uses an ACT tapped delayline to accomplish the delay and sensing operations required by thetransversal filter. The non-destructive sensing electrodes T sense theinput signal as it propagates down the delay line D. Each sensing tap Tis connected to a common summing bus B (the accumulator of thetransversal filter) through an individual programmable attenuator. Thevalues of the attenuators (the weighting coefficients) are set by adigital controller. All tap weights W are stored in random access memory(RAM which is monolithically integrated with the ACT delay line.

An ACT programmable transversal filter has 128 taps that may be set toany of 31 values between +1 and -1 (5-bit tap weighting). The inputsampling rate of the device is 360 MHz, and the center to center tapspacing is 5.6 ns. This gives a single-tap bandwidth of 180 MHz, and afilter Nyquist interval of 90 MHz. A single tap may be programmed inless than 1 μs, and the entire device may be programmed in under 100 μs.

FIG. 48 illustrates a notch filter utilized in the IBOC FM-DAB receiveras the tracking notch 150 of FIG. 11. Notch filter N has an ACTtransport channel 1900, with SAW generation, input and output pursuantto U.S. Pat. No. 4,633,285. Three groups of non-destructive sense (NDS)electrodes or taps are spaced along the surface of channel 1900 forsensing the charge packets moving through the channel with the SAW.

NDS taps 1902, 1904, 1906, and 1908 are positioned at the input end ofchannel 1900 and are the reference taps. NDS taps 1910, 1912, 1914, and1916 are at the output end of channel 1900 and are the delay taps. NDStaps 1918, 1920, and 1922 are disposed between the reference and delaytap groups and are used to provide a quadrature waveform for use by thephase detector.

Buffer circuits 1924, more particularly explained later, are connectedto the output of the taps of the reference, delay, and quadraturegroups. Taps 1904, 1906 of the reference group are static, as are taps1912, 1914 of the delay group. Taps 1902 and 1904 are dynamic andadjustable as a result of tap weight control voltages A and A',respectively. Similarly, taps 1910 and 1916 are dynamic and adjustableas a result of tap weight control voltages A' and A, respectively.

Buffer circuits 1926, which correspond to buffer circuits 1924, areconnected to outputs of the weighting circuits of the adjustable taps1902, 1908, and 1910, 1916. Summers 1928, 1930, and 1932 are thenconnected to the outputs from the static and dynamic taps of thereference, delay, and quadrature tap groups, respectively.

The reference and delay tap groups are spaced 700 nanoseconds apart inan ACT channel. Each of these tap groups consist of four buffered NDSelectrodes. The center two electrodes in each of these tap groups arethe static taps and the outside electrodes are the dynamic taps.Weighting circuits in the dynamic taps--as well as buffer circuits forall taps--are taken from FIG. 33. The dynamic taps in the reference anddelay groups cause the relative group delay between their summed outputsto increase and decrease. An additional quadrature tap group consistingof three unweighted electrodes is used to provide a quadrature waveformfor use by the phase detector.

FIG. 31 describes three different FIR filter variable delay elementswith different sensitivities to the dynamic tap weights. FIG. 31illustrates how the structure of the background taps 510, 514, 516, 518and 526 affects the sensitivity of relative tap group delay to theweight A of the dynamic taps 512, 520, 522 and 524. In each case shownin FIG. 31, the dynamic tap weight A is controlled by a five bit binaryword (one sign bit, 4 magnitude bits). A "delay increment" correspondsto a one least significant bit (LSB) change in the five bit amplitude A.These examples are based on transversal filter tap spacing of 5.6nanoseconds. The results given are valid near 10.7 megahertz.

Referring to FIG. 31a, in the first case 528, a large number of taps 510and 514 having static weights yields a large inertia of group delay. Aone LSB change in the weight A of the dynamic taps 512 yields only a 20picosecond change in the group delay of the delay element 528. In thesecond case 530, a smaller number of static taps 516 and 518 allows thedynamic taps 520 to exercise more significant control over the relativegroup delay of the delay element 530. In this case, one LSB of change inthe weight A of the dynamic taps 520 yields a 100 picosecond change inthe delay element group delay. Referring to FIG. 31c, in the last case532, a relatively small number of static taps 526 and a wide spacing ofthe dynamic taps 522 and 524 yields a very wide variation in group delayfor the delay element 532 with respect to the weight A of the dynamictaps 522 and 524. A one LSB change in A in case 532 yields a 500picosecond increment in relative group delay for the tap group.

When a pair of dynamic taps are changed to adjust the relative delay ofa tap group, the magnitude of that tap group also changes. This shouldbe avoided because deep cancellation in a simple delay element notchfilter requires that the amplitude of the delayed signal 444, 452 matchvery closely with the amplitude of the reference signal 442, 450. FIG.32 illustrates a structure which compensates for these undesiredmagnitude variations. In FIG. 32, both the reference 540 and the delay542 elements are implemented by transversal filter variable delayelements. The reference and delay variable delay groups 540 and 542 aremirror images and are controlled symmetrically. As A is raised, thereference group dynamic taps 548 and 550 cause the reference group toyield lower relative delay, while the delay group dynamic taps 552 and554 cause the delay group to yield higher relative delay. The delaybetween the reference 540 and delay 542 groups increases, but themagnitudes of the groups 540 and 542 track each other. Similarly, as Ais lowered, the delay between the reference 540 and delay 542 groups islowered, but again the magnitude of the delay group 542 tracks themagnitude of the reference group 540. Cancellation depth at 448 ismaintained because the magnitudes track. The tap groups 556 and 558 inFIG. 32 are used to generate signals whose interference components areroughly in quadrature with the interference components at the referencetap group 540 and at the delay tap group 542. These approximatequadrature signals are used in the phase detector of FIG. 35.

Circuits used to tap off a tapped delay line are shown in FIGS. 33a and33b. These circuits have been combined to implement the variable delayelements shown in FIG. 32 and used for setting the tap weights in FIG.48. The buffer 560 corresponds to buffer circuits 1924 and 1926 and isused to tap off a delayed signal from an analog tapped delay linewithout loading down the delay line. The analog weighting circuit 562 isused to implement a variable tap weight A of taps 1902, 1908, and 1910,1916 of FIG. 48. The analog weighting circuit 562 requires twocomplementary control voltages, A 564 and A' 566, to implement anattenuation proportional to A through a FET voltage divider. Theweighting circuit 562 is typically followed by a buffer similar to thebuffer 560, so that the tap weighting is insensitive to output loading.When this circuit is applied to the types of tap groups shown in FIGS.32 and 30, tap group relative delay ΔT is very linear with controlvoltage A.

The circuits shown in FIGS. 33a, and 33b may be used to implement analogtap weighting on the dynamic taps in each tap group. The buffer 560circuit shown is a source follower. The buffer is used to sense thevoltage of a nondestructive sense (NDS) tap in the ACT channel whoseposition corresponds to the intended location of a dynamic tap. The NDStap conveys a high impedance voltage signal in proportion to theproximate charge sensed in the ACT channel. The buffer circuit sensesthis voltage with a high impedance input (V_(in)) and supplies theweighting circuit 562 with that same voltage (V_(out)) but at a lowimpedance suitable for driving the weighting circuit.

The weighting circuit 562 is a resistive voltage divider whose resistiveelements are field effect transistors. The control voltages A 564 and A'566 control the effective resistance of the FETs in the voltagedividers. Depletion mode FETs are used which are compatible with the ACTfabrication process. The result is that the effective attenuation fromV_(in) to V_(out) in the weighting circuit 562 may be varied linearlyabout 0.5 by varying A and A' linearly, in a complementary fashion,about -1.25 V with respect to ground as shown in the weighting circuit562.

Operation of the weighting circuit 562 is sensitive to impedance loadingat V_(out). For this reason, the weighting circuit is followed byanother source follower buffer 560 circuit. This last buffer circuitdecouples the weighting circuit from its eventual load.

Successful design of a delay element tap group involves assuringsufficient group delay variation to accomodate sufficient range ofcontrol over notch frequency. The required one-sided group delayvariation, ΔT, is related to the one-sided peak frequency deviation bythe following equation ##EQU5## where f_(c) is the center operatingfrequency and T_(c) is the nominal baseline delay. In the preferredembodiment, f_(c) =10.7 MHz, T_(c) =701 ns and Δf=100 kHz. ΔT istherefore greater than or equal to 7 ns to provide for group delayvariation which will yield sufficient (100 kHz) notch frequency range.

It should be noted that the use of complementary tap groups, such asthose shown in FIG. 32, reduces by a factor of 2 the required groupdelay variation range of each tap group. This is because the relativetap group group delay changes as the sum of the changes in the twoindividual tap group group delays. Therefore, in the preferredembodiment, since complementary tap groups are used, the required groupdelay variation per tap group is only 3.5 ns.

The maximum phase deviation, ΔΦ, required of each tap group may beapproximated from the maximum group delay variation, ΔT, required ofeach tap group, using the equation

    ΔΦ=2πf.sub.c ΔT=2πT.sub.c Δf   (13)

For the preferred embodiment, ΔΦ=0.24 Rad=14°.

The range of amplitude variation range of the dynamic taps with respectto the amplitudes of the static taps should be sufficient to accomodatea phase variation of ΔΦ. The phase variation for a given tap groupstructure of static and dynamic taps may be analyzed as follows.

The baseline magnitude of a tap group may be taken by analyzing theresponse of a tap group at center frequency. Suppose a tap groupconsists of N static taps of unit amplitude spaced at intervals T_(s)surrounded by a pair of dynamic taps, one at each end. The baselinemagnitude response at center frequency may be found by evaluating theFourier transform of the tap group with the dynamic taps set to zero.The magnitude M of the resulting frequency response is ##EQU6## If thedynamic taps are set to zero, the phase response of the tap group iszero with respect to the center of the tap group. If the dynamic tapsare set to values of ±A, then the dynamic taps add an odd symmetrycomponent of of magnitude M_(odd) which is

    M.sub.odd =2A sin (π(N+1)f.sub.c T.sub.g)               (15)

Sufficient phase deviation range is assured when A may be varied suchthat ##EQU7## In the preferred embodiment, T_(s) =16.67 ns, N=2, M=1.6and M_(odd) =0.4 for A=0.2. This means that a dynamic tap weight ofA=0.2 will a phase offset of ΔΦ=14° which will adjust the notchfrequency by 100 kHz in the preferred embodiment.

A block diagram for a comprehensive variable delay element cancelercircuit is shown in FIG. 34. This integrated canceler circuit accepts,as an input, an IF with an interference signal 570. Reference 544,quadratures 556, 558 and delay 546 tap groups are of the formillustrated in FIG. 32 as 544, 556, 558 and 546 All tap groups tap offthe same tapped delay line. Any tapped delay line technology may beemployed. Control is applied to the reference group through the leftcontrol line 580. Control is applied to the delay group through theright control line 582. The center control lines 584 control the gainadjustments 586 and 588 for the canceler summation circuit 446. Theability to directly control the attenuation 586 and 588 into the summer446 allows for calibration of null depth at the output of the canceler448. The cancelled signal 448 is an output of the integrated cancelercircuit. The signal coming from the reference tap group 594, thequadrature tap groups 596 and 598, and the delay tap group 600 areoutputs which are used as inputs to the phase detector.

Phase Detector

FIG. 35 shows the phase detector used in tracking the interferencefrequency. R 594, reference signal, is the signal taken from thereference leg of the integrated canceler circuit of FIG. 34. D 600,delayed reference, is the signal taken from the delay leg of theintegrated canceler circuit. Q 596 and Q 598, approximate quadraturesignals, are taken from the quadrature outputs 556 and 558 of theintegrated canceler circuit.

The first function applied to the phase detector inputs is zero crossingdetection (one bit D/A conversion) at the zero crossing detector 618.The predominant interfering signal is expected to dominate the locationsof the zero crossings. The outputs 620, 622, 624 and 626 of the zerocrossing detector are square wave estimates of the predominantinterfering component. These square wave estimates are referred to bythe same names R, D, Q and Q as their corresponding zero crossingdetector inputs 594, 600, 596 and 598.

The two level digital (square wave) estimates of the interference signalare applied to two AND gates. The first AND gate is the phase leaddetection gate 628, and its output is denoted RDQ 632. The second is thephase lag detection gate 630, and its output is denoted RDQ 634. Theoperation of these gates is illustrated in FIG. 36.

FIG. 36a shows the desired condition of D 622 equal to the inverse of R620. This condition yields maximum cancellation of the interferencesignal at the canceler output 448. In this case, the phase leaddetection signal RDQ 632 and the phase lag detection signal RDQ 634 areboth equal to zero at all times.

FIG. 36b shows D 622 leading R 620 in phase. This case indicates thatthe notch frequency f_(z) 470 is higher than the frequency of theinterference signal. The delay between the reference group and the delaygroup is too short for optimal cancellation. This condition is indicatedby the presence of a pulse on the RDQ line 632 and the absence of apulse on the RDQ line 634. The necessary correction to the controlvoltage A 548, 554, 564 is proportional to the width of the pulse on theRDQ line 632.

FIG. 36c shows D 622 lagging R 620 in phase. This case indicates thatthe notch frequency f_(z) 470 is lower than the frequency of theinterference signal. The delay between the reference group and the delaygroup is too long for optimal cancellation. This condition is indicatedby the presence of a pulse on the RDQ line 634 and the absence of apulse on the RDQ line 632. The necessary correction to the controlvoltage A 548, 554, 564 is proportional to the width of the pulse on theRDQ line 634.

FIG. 37 shows how the control line signals 548, 554, 564, 580 and 582are derived from the phase detector output signals RDQ 632 and RDQ 634.The phase lead and phase lag signals RDQ 632 and RDQ 634 are subtractedin a summer 640 and then integrated in an analog integrator 642. Theoutputs of integrator 642 are added in an operational amplifier circuitsB so that the outputs 580 and 582 therefrom may generate the controlvoltages for tap weights A and A'. Amplifier B operates so that A=-1.25V+B and A'=-1.25 V-B, where B equals the output from integrator 642. Theresults 580, 582 are used to generate the analog control voltages A 548,554 and 1-A 550, 552. These voltages in turn affect the delay T 440between the reference 544 and delay 546 tap groups to close the feedbackloop on tracking cancellation.

Closed Loop Canceler

FIG. 38 is a block diagram of a closed loop tracking delay element notchfilter. The input 570 is the signal of interest plus one predominantinterfering signal. The adjustable delays, implemented using tappeddelay line 652, are the reference elements 594, 544 and the delayelements 600, 546. The delay generator 652 also implements stationaryquadrature delay elements 596, 598, 556, and, 558. The reference 594,544 and the delay elements 600, 546 are summed at the summer 446 toyield the output signal 448 with interference cancelled.

The phase detector and control signal generation circuitry shown inFIGS. 35 and 37 are implemented in phase detector 664. Its output 668dynamically controls the adjustable delay line 652. Output 668 alsoserves as an estimator of the interference frequency and may be used togenerate a demodulated FM output. The clock 666 is not essential, but isused in a digital implementation to be discussed later. The FIG. 38tracking canceler may be implemented using either digital or analogdelay control.

A tracking canceler loop employing analog delay control is illustratedin FIG. 39. The reference 544 and delay element 546 wave forms aresummed at 446 to yield a cancelled interference output 448. Gainadjustment elements 586 and 588 ensure deep cancellation, as shown inFIG. 34. The signals R 594, Q 596, Q 598 and D 600 are applied to levelthresholding circuitry 618 which is followed by high speed logic 628,630. Elements 618, 628, 630 represent the phase detector of FIG. 35.High speed logic 628, 630 is followed by the analog integration function640, 642 which is shown in FIG. 37. This yields analog control signals580, 582 which control the adjustable delay elements, and a demodulatedFM signal in the case where FM demodulation is useful.

A tracking canceler loop employing digital control is illustrated inFIG. 40. The FIG. 40 canceler is essentially the same as the FIG. 39canceler. The main difference is that the analog integration elements640, 642 of FIG. 39 are replaced by an edge detector 680 and a digitalup-down counter 682. These are used to count phase detector pulses 632,634 as a means for integrating numerically. The output of the digitalcounter is interfaced to a set of digital control lines 684 which areused to digitally set the weights of the dynamic taps 548, 550, 552 and554 in the reference 544 and delay 546 elements. The digital count in682 may also be passed through a digital to analog (D/A) converter 686to yield an estimate of the interference frequency 688.

The notch filter of FIG. 26 may be modified so that one input to thesummer 442 is negative and the other positive. The wideband frequencyresponse would be defined using

    |H.sub.i (f)|=|sin(πfT)|(17)

The zeroes would be defined using ##EQU8## And the narrowband modelwould reflect these new cancellation frequencies.

Another variation on delay element cancellation is the case where one ofthe legs 442 or 444 into the summer 446 includes a bandpass filter forcoarse isolation of the interference signal. In this case, bandwidths ofoperation in excess of 1/T are possible.

Plural tracking cancellers may be cascaded for increased cancellationdepth. It is likely that the cancellation frequency of the cancellerswould be established by a single phase detector at the first canceller.

Another variation may be realized through a compound approach involvingcancellation through the weighted summation of combinations of staticand adjustable tap groups.

The fundamental building block of a delay element canceller is a delayelement 440. More precisely, any tapped delay line 490 may be used toimplement the delay elements 544, 556, 558 and 546 needed for a trackingdelay element canceller. An acoustic charge transport tapped delay lineis preferred. Tapped delay line implementations which could be adaptedinclude: digital signal processing components (DSP's); bucket brigadedevices (BBD's); charge coupled devices (CCD's); sets of A/D convertersfollowed by banks of shift registers and D/A converters (A/D, digitaldelays, D/A); delta-sigma modulators followed by a long shift registerand banks of summers (A/D, digital delay, D/A); and, transmission (L-Cor R-C delay) lines.

Additionally, a reactive component delay element may be used to adjustdelay monotonically with a control signal. The adjustable reactive delayelement may be used in a canceller by itself, in cascade with a delayline of constant delay, in cascade with one or more reactive delayelements of constant or variable delay, or in cascade with a delay lineelement whose delay is also adjustable. Furthermore, adjustable reactivedelay elements could be used in one leg 442, 444 of the canceller goingto the summer 446, while an adjustable delay line delay could be used inthe other. Still another variation is to combine bandpass filtering,mentioned as a vehicle for coarse interference isolation, with theseapproaches. It is conceivable that the reactive element used to adjustdelay could be one or more of the components of the same coarse bandpassfilter or filters.

FIG. 31 illustrates three variations of adjustable delay tap groupswhose dynamic taps are of opposite sign (FIG. 30c). These threevariations are intended as examples. The same methodology may be appliedto any number of tap groups in either the opposite sign dynamic tapscase (FIG. 30c) or in the complimentary positive (or negative) dynamictaps case (FIG. 30b).

Another delay element tap group is illustrated in FIG. 41. In this tapgroup, the elements achieve a wider range of delay adjustment by"walking" along the tapped delay line. A pair of dynamic taps 702, 704in FIG. 41a are adjusted until they reach their adjustment threshold,while taps 700 are static. At this point, the dynamic taps "walk" over706, 708 in FIG. 41b to a new position where they resume adjusting thedelay 712, 714 in FIG. 41(c). In that case, taps 710 are static. Thistype of tap walking or transition may occur more than once and may occurin either direction. This walking delay element tap group also lendsitself to symmetric reference and delay structures such as that shown inFIG. 32.

In FIG. 32, either the reference group 544 or the delay group 546 couldbe replaced by a stationary tap group. In this way, system simplicity isgained at the expense of cancellation depth. Furthermore, the 90° (556)and -90° (558) delay groups could come from a single tap and aninverting amplifier circuit, or the ±90° signals could be generatedusing reactive elements. The precise phase of these circuits is notcritical to the operation of the phase detector described.

The tap weight structures mentioned for controlling dynamic tap weightsinclude an analog FET voltage divider structure (FIG. 33a) or a digitalcontrol structure inherent to the ACT PTF (FIG. 33b). Any tap weightimplementation which would control the tap weight in real time could bea suitable approach.

The phase detector of FIG. 35 provides a convenient indicator of theoffset between the instantaneous loop cancellation frequency and theinterference frequency. However, the use of this specific phase detectoris not absolutely necessary. An alternative would be to use either a180° or a 0° phase phase/frequency comparator, such as the GigaBit Logic16G044/16G044M. An advantage of such a phase/frequency detector is thatthe Q 596 and Q 598 outputs are not needed for the phase detector.

Cancellation may be configured such that a master canceller receives asignal from an antenna pointed to the interference, while a second slavecanceller receives its input from a second antenna pointed to the sourceof interest. The master canceller has a high interference to signalratio which makes it easier for it to lock onto the interferencefrequency, while the slave canceller, which may not have enoughinterference to lock onto, yields the more desireable processing gain ofthe canceller compounded with processing gain due to directivity.

For FM demodulation applications, demodulation quality is superior toPLL demodulation in environments with significant multipath. This isbecause the canceller tracks frequency rather than phase. The cancelleris therefore insensitive to abrupt phase transitions which would cause aPLL to lose lock.

When the dynamic tap weight A 548, 554 is used to demodulate an FM inputsignal, the demodulation linearity calculation is straightforward andcompensation for nonlinearity is also straightforward. Suppose, however,the FIG. 33 tap weight circuits were employed, and the FET gate voltageA 564 were used as the demodulation output. In this case, the dividernetwork and the components themselves compound the demodulationnonlinearities. Linearization circuits are still practical in this case.Linearization of the direct demodulated signal represents an importantvariation of this invention for FM demodulation applications.

Nonlinearities and compensation for the simple case of demodulationthrough direct tap weight A 548, 554 is now considered as an example.Suppose the R 544 and D 546 tap groups were separated in relative groupdelay by T 440 to yield a given cancellation frequency f where.

    T=T.sub.c +ΔT                                        (19)

and

    f=f.sub.c +Δf                                        (20)

T_(c) represents the nominal delay when A takes on its nominal averagevalue (1/2 in 500 or 0 in 506) which corresponds to the centercancellation frequency f_(c). Δf and ΔT represent relative cancellationfrequencies and delays. ##EQU9## may be rewritten ##EQU10##

The variable N represents n+1/2 when the canceller summation 446operates on R 544 and D 546 with the same sign. The variable Nrepresents n when the canceller summation 446 operates on R 544 and D546 with opposite sign. For a given applied FM signal with instantaneousrelative frequency Δf, the phase Φ(Δf) relative to the center of avariable tap group is denoted ##EQU11## N_(var) is an index whereN_(var) =2 if 2 variable delay groups are combined in a complementarydelay structure (FIG. 32) and N_(var) =1 if only one tap groupimplements a variable delay. Since ΔT may be rewritten ##EQU12## Φ(Δf)may also be rewritten ##EQU13##

The phase relative to the center of a variable tap group is also the arctangent of the ratios of the odd and even components of the timeresponse of a tap group. The even component of a tap group's timeresponse is determined by the static taps 498, 504 and the averagevalues of the dynamic taps 500, 502, 506 and 508. This component may bedenoted M_(even). M_(even) is constant for a given tap group. The oddcomponent of a tap group's time response is determined by the dynamictaps and may be denoted M_(odd). M_(odd) is a linear function of A 548,554 and may therefore be used to characterize nonlinearities of A withΔf. The mathematical result is that ##EQU14## Equating Φ(Δf) in Eqns. 17and 18 yields ##EQU15## which describes the nonlinearity of A (M_(odd)(Δf)) with respect to Δf. Eqn. 24 characterizes nonlinearities due to FMdemodulation. As an example, suppose a sinusoid was used to frequencymodulate a 10.7 MHz carrier (f_(c) =10.7 MHz) with a maximum frequencydeviation of 85 kHz. Suppose also that the tracking cancellerdemodulated incorporated a nominal delay of 701 ns (T_(c) =7.5/f_(c)=701 ns) in a complementary variable tap configuration (N_(var) =2). Inthis case, Eqn. 24 predicts a total harmonic distortion of 0.23%.

A simple linearization algorithm to improve this would be to applyhyperbolic tangent compression. Bipolar transistor differential paircircuits have hyperbolic tangent gain characteristics. The compensationnonlinearity is modeled mathematically using ##EQU16## where M_(comp)(Δf) represents the compensated demodulation waveform. Applying the sameexample, total harmonic distortion is reduced from 0.23% to 0.0016%.This represents a very significant improvement in FM demodulationlinearity through a very simple linearization circuit.

The tracking delay element interference canceller and FM demodulator maybe modified to make a simple FM demodulator. An example of thismodification is shown in FIG. 42. The incoming FM signal 750 is passedthrough a delay network, containing a reference element 752 and a delayelement 754. In this simple case, the relative delay between thereference 752 and delay 754 elements tracks the instantaneous FM signal750 by maintaining a 90° phase relationship between the two timingelements 752 and 754. Control of the variable group delay elements isthrough the control line 764. FIG. 42 shows that the delay element 754is variable, while the reference element 752 may not be. In practice,either or both may be made variable.

Instantaneous phase relative to the 90° baseline reference is measuredthrough a mixer 756 and a lowpass filter 758. The integrator 760accumulates the error signals generated by the phase detector 756, 758and passes the result to the tap control block 762 which generates theproper control signals to close the feedback loop. The optionallinearization section 766 may be used to compensate for nonlinearitiesin the tap control or in any aspect of the loop. The final demodulatedFM output 768 is the result.

In cases where the interferer becomes temporarily weak, undesiredspurious signals may dominate the phase detector RDQ 632 and RDQ 634lines. When passed through the integrator 642 or counter 682 of FIGS. 39and 40, respectively, the result is an error in the control signals 580,582 of the delay elements 544, 546, 754. The remedy is to sense thelevel of the interference at the input. When the interference signal 570power is sufficient to drive the phase detector 664, the integrator 642or counter 682 is enabled. When the interference signal 570 power isinsufficient to drive the phase detector 664, the integrator 642 orcounter 682 is disabled. When the integrator 642 or counter 682 isdisabled, the control lines 580, 582 of FIG. 37 remain at their lastknown states until interference power is sufficient to enable properphase detector operation. This variation improves tracking performancefor FM demodulation applications as well as interference cancellationapplications.

Adjustable delay line element cancellation offers the advantage of fastresponse to delay control over systems using adjustable reactive elementcancellation because reactive elements ring. This adjustment is fasterthan systems which use numerical interference frequency estimationtechniques, because the frequency tracking is relatively trivialcompared to conventional interference frequency estimation methods. Thesimple phase detection method works because the same signals that canceleach other are used to track each other. Quick control makes thiscanceller suitable for fast tracking of the optimal cancellationfrequency. Another advantage of the delay element canceller over areactive element notch filter is that when tracking quickly, thepositive and negative phases of the skirts in the cancellation frequencyresponse causes residual high frequency BPSK conversion on theinterference signal which yields a higher effective cancellation depth.A third advantage of the delay element canceller is that the gradualskirts of the canceller frequency response make it slightly lesssensitive to small errors in cancellation frequency than a reactivenotch canceller with steeper skirts.

The delay element canceller has linear skirts, rather than thearbitrarily steep skirts of a reactive element canceller. This couldcause unwanted distortion in the desired signal. Another cause ofundesired distortion is the opposing phase characteristic of thecanceller frequency response. Opposing phases on either side of thecancellation frequency helps cancellation but could hurt the desiredsignal. Solutions which correct the opposing phases of opposingcanceller frequency response include the cascaded canceller and compoundcanceller approaches mentioned above. Unfortunately, these solutionsalso degrade the slopes of these same skirts.

In terms of the preferred phase detection method (FIGS. 35-37) theadvantages include implementation simplicity, simple direct control ofphase detector sensitivity through loop dynamics, the ability to detectnew relative phase information every carrier cycle and the availabilityof simple enable and disable control. A disadvantage of the phasedetector is that it requires an approximate quadrature (˜90°) waveform,and that its operating range is limited by the tracking accuracy of this90° offset.

In terms of FM demodulation, one advantage over PLL's is that loss oflock due to phase reversals is avoided. Loss of lock yields a severeundesired transient in a PLL demodulation waveform, while it only yieldsa mild temporary error in the DLL demodulator described. Anotheradvantage over conventional PLL demodulation of FM is that a VCO is notrequired. A disadvantage is that a delay element is required. Anotherdisadvantage to PLL's is noise performance. Tracking a signal withitself through a delay yields better resistance to artificial phasetransitions (multipath). Tracking a signal with a VCO alleviates a noisesource (the delayed input signal). This is interpreted to mean thatmultipath resistance is bought at the expense of signal to noise ratioin clear channels.

A simple variable delay line approach yields an easily frequencyadjustable signal canceller. Simple control of delay line cancellerallows direct linear control of cancellation frequency. The inherentabsence of resonant elements allows for fast adjustment of the frequencyof cancellation. A single pair of gates serves as a phase detector for afast, easy and simple way to track a single predominant interferer. Theresulting closed loop system is a self adjusting, fast trackinginterference signal canceller.

In this fast tracking loop, the loop delay is locked in delay to thecorresponding frequency of the predominant input signal. Precisetracking of the frequency of the predominant input signal allowsfrequency tracking and FM demodulation. When very linear FM demodulationis required, linearization circuits may easily be used to improvedemodulation linearity.

The delay line structure is inherently simple. The cancellation processhas a very short time impulse response length. The canceller's responseto the control signal is very fast because of the absence of resonantelements in the canceller. The canceller may be disabled very easily byturning off one of the inputs to the signal combiner.

The phase detector allows for fast tracking of the relative phase ofinterfering signals within the operating range. The sensitivity of thetracking canceller to the input is easily adjusted in the phase detectorby adjusting the sensitivity of the zero crossing detectors 618. Thesensitivity of the canceller may also be easily adjusted through thefeedback network which controls the tap weights. The signals whichdetermine the cancellation phase at the signal combiner 446 are the samesignals that are used to update the cancellation phase at the signalcombiner. This assures cancellation tracking. The lack of resonantelements in the control of this update (i.e. the delay, which controlscancellation frequency) enables true cancellation frequency readjustmenton a cycle by cycle basis of the interfering signal.

This is not a frequency domain canceller nor a time domain canceller,but a variable notch whose instantaneous null frequency is automaticallyadjusted in real time. The multipath resistant nature of resulting FMdemodulation is due to an adjustable delay which tracks instantaneousfrequency rather than an adjustible oscillator which tracks thesecondary variable, instantaneous phase.

The primary effects of multipath in an IBOC DAB system and thetechniques used to mitigate these effects are summarized in Table 1.Inter-symbol interference is caused by strong echoes with long delaytimes; most experimental evidence suggests that delay spreads of 1 to 5microseconds are common, although some measurements indicate that delaysof 15 or even 30 microseconds may occur under some circumstances. Asdescribed above, one common technique used to combat this effect isfrequency-domain multiplexing (FDM), in which a high data-rate signal isdivided into a number of lower data-rate signals which are thentransmitted in lower-bandwidth subchannels. The symbol time in eachsubchannel is made longer than the longest expected delay; this suggestssubchannel data rates of less than 33 kHz (1/30 microsec) for theconditions described above.

FDM has the effect of confining the impact of long-delay multipath to asmall number of subchannels; combining FDM with frequency sliding andforward error correction provides significant system robustness againstmultipath. The primary benefit of

                  TABLE 1                                                         ______________________________________                                        Multipath         Mitigation Technique                                        ______________________________________                                        Inter-symbol interference                                                                       Frequency-Domain MUXing                                     (caused by strong Frequency Sliding                                           echoes with long delays)                                                                        Forward error correction                                                      Channel equalization                                        Amplitude fading  Frequency Diversity                                         (bandwidth inversely                                                          Subchannel spreading                                                          proportional to delay                                                         Frequency sliding                                                             spread)                                                                       Coding and interleaving                                                       ______________________________________                                    

frequency sliding in this regard is to minimize the effect of multipathon any one channel by moving the subchannels through the affectedfrequency range. Forward error correction may then be applied to restorethe data lost in the affected subchannels. Additional protection againstinter-symbol interference is provided by the channel equalizationtechniques described above.

These techniques offer some amount of protection against the amplitudefading effects of multipath, although the amount of protection dependson the characteristics of the fading. In particular, the coherencebandwidth of the fading is a critical factor in determining theeffectiveness of these mitigation techniques. Experiments have suggestedthat typical rural and urban multipath will cause fading with coherencebandwidths of

    B=30/D to 60/D

where B is the 90% coherence bandwidth in kHz and D is the delay spreadin microseconds. For delay spreads of more than 1 microsecond, thisgives coherence bandwidths of less than 30 kHz. In these cases,amplitude fading will affect only a small number of subchannels, and thefrequency diversity techniques described above will be effective.

For cases with very small delay spreads (and correspondingly largecoherence bandwidths), multipath mitigation through data coding,interleaving, and error correction are required. For mobile receivers,the spatial correlation of amplitude fading and the velocity of theplatform are important, since these will determine the amount of timethe receiving antenna spends in the fading region. Measurements indicatethat spatial correlation distances from 7 inches to 2 feet are common;theoretical analysis predicts correlation distances of up to 7 feet. Fora vehicle travelling at 20 miles per hour, these multipath correlationdistances would require data coding and correction which can handlereduced-quality data for periods of tens to hundreds of milliseconds.This is within the capabilities of recently developed burst-errordetection and correction techniques.

FM TO AM CONVERSION CANCELER

In-band on-channel FM DAB requires demodulation of a low power digitalsignaling waveform in the presence of a high power FM signalingwaveform. A delay locked loop FM canceler has been proposed; however,multipath has the effect of causing conversion from FM to AM. Theresulting undesired AM signal interferes with the DAB but is notcompletely extracted with a delay locked loop canceler or other known orproposed cancellation techniques.

The FM to AM conversion canceler functions upon the principle that theamplitude of the FM to AM conversion interference component is correctedto the instantaneous frequency of the FM signal as a function of thechannel multipath. The FM to AM conversion canceler estimates thiscorrection and continuously updates this estimate. While continuouslyupdating this correlation estimate, it uses the instantaneous frequencyand the correlation estimate to estimate the FM to AM interference termfor cancellation.

A lookup table embodiment is shown in FIG. 45. The FM canceler 902receives FM DAB signal 900 and provides a baseband canceled signal (DABminus FM) amplitude as well as instantaneous frequency information to"lookup table generation" hardware 908 used to generate a "lookup table"912--a running estimate of the FM to AM conversion interference to beexpected from a given instantaneous FM frequency. The lookup tablereceives the instantaneous frequency and returns the most recentestimate 916 of the FM to AM interference level to be expected at thatfrequency. This FM to AM interference estimate is subtracted from theDAB composite to yield a clean DAB composite 924, free of most of its FMto AM interference.

A second embodiment based on polynomial channel estimation is shown inFIG. 46. In this case, the FM signal frequency estimate drives apolynomial generator which estimates the FM to AM conversioninterference 954 in the channel. This estimate is subtracted at 938 fromthe raw DAB composite 936. The coefficients 970 of the polynomial arederived by measuring and integrating the cross-correlation between theresulting DAB composite 960 and each term of the polynomial. In thismanner, the coefficients of the FM to AM cancellation polynomial arealways adjusted so as to drive the FM to AM interference terms to zeroafter cancellation.

A third embodiment, based also on polynomial channel estimation, may beused for systems where DAB processing takes place at an intermediatefrequency (IF). This system is illustrated in FIG. 49. Two separatepolynomials are used to continuously estimate and cancel the in phaseand quadrature components of the FM to AM interference.

While this invention has been described as having a preferred design, itis understood that it is capable of further modifications, uses and/oradoptions of the invention following in general the principle of theinvention and including such departures from the present disclosure ascome within the known or customary practice in the art to which theinvention pertains and as may be applied to the central featureshereinbefore set forth, and fall within the scope of the invention andof the limits of the appended claims.

We claim:
 1. A method for broadcasting information in a composite signalfrom a transmitter to one or more receivers, comprising:a) frequencymodulating an FM message to generate an FM waveform; b) multiplying saidFM waveform by an AM message to generate a suppressed carrier AM over FMsignal; c) combining said suppressed carrier AM over FM signal with saidFM waveform to generate a large carrier AM over FM composite signal; d)transmitting said large carrier AM over FM composite signal; e)receiving said large carrier AM over FM composite signal; f) recoveringone of said FM message and an AM waveform corresponding to said AMmessage over FM composite signal.
 2. The method of claim 1 wherein therecovering step comprises the steps of selecting one of said AM and FMwaveforms from said large carrier AM over FM composite signal anddemodulating the selected waveform.
 3. The method of claim 1 wherein therecovering step comprises the steps of:a) isolating said FM waveformfrom said large carrier AM over FM composite signal to generate an FMonly signal; and b) demodulating said FM only signal to recover said FMmessage.
 4. The method of claim 1 wherein the recovering step comprisesthe steps of:a) isolating said AM message from said large carrier AMover FM composite signal to generate an AM only signal; and b)demodulating said AM only signal to recover said AM message.
 5. Themethod of claim 1, further comprising the steps of:a) emphasizing a rawAM message to generate said AM message, for subsequent use inmultiplying said FM waveform; and b) deemphasizing said recovered AMmessage with deemphasis filter to generate an AM message afterdeemphasis.
 6. A method for transmitting information, comprising thesteps of:a) frequency modulating an FM message to generate an FMwaveform; b) multiplying said FM waveform by an AM message to generate asuppressed carrier AM over FM signal; c) summing said suppressed carrierAM over FM signal with said FM waveform to generate a large carrier AMover FM signal; and d) transmitting said large carrier AM over FMsignal.
 7. A method for transmitting information, comprising:a)frequency modulating an FM message to generate an FM waveform; b)multiplying an AM message with a local oscillator signal to generate adouble sideband suppressed carrier AM signal; c) summing the localoscillator signal and the double sideband suppressed carrier AM signalto generate a double sideband large carrier AM signal; d) multiplyingthe FM waveform with the local oscillator signal to generate a firstintermediate signal; e) band-pass filtering the first intermediatesignal to produce an FM waveform at sum frequency signal; f) multiplyingthe FM waveform at sum frequency signal with the double sideband largecarrier AM signal to produce a second intermediate signal; g) band-passfiltering the second intermediate signal to produce a double sidebandlarge carrier AM over FM signal; and h) transmitting said doublesideband large carrier AM over FM signal.
 8. A method for recoveringsignals from a transmitted large carrier AM over FM signal incorporatingan AM message and an FM message, comprising:a) receiving the largecarrier AM over FM signal; b) obtaining an FM only signal; c)multiplying said FM only signal with a local oscillator signal togenerate a first product signal; d) generating from the first productsignal a high IF FM signal; e) combining the high IF FM signal with thelarge carrier AM over FM signal to produce a second product signal; f)generating from the second product signal an AM only signal; and g)demodulating said AM only signal to recover said AM message.
 9. Themethod of claim 8 wherein the step of generating from the first productsignal an FM at high IF signal comprises band pass filtering said firstproduct signal.
 10. The method of claim 8 where the step of combiningthe FM at high IF signal with the large carrier AM over FM signal toproduce a second product signal comprises multiplying the FM at high IFsignal with the large carrier AM over FM signal.
 11. The method of claim8 wherein the step of generating from the second product signal an AMonly signal comprises band-pass filtering the second product signal. 12.The method of claim 8 further including cancellation of crossinterference.
 13. A method for recovering signals from a transmittedlarge carrier AM over FM signal incorporating an AM message and an FMmessage, comprising:a) receiving the large carrier AM over FM signal; b)generating, from said large carrier AM over FM signal, a suppressedcarrier AM over FM signal; c) generating, from said large carrier AMover FM signal, an FM only waveform; and d) demodulating the largecarrier AM over FM signal, wherein said demodulating step comprisescombining said suppressed carrier AM over FM signal and said FM onlywaveform to produce a product signal and generating from said productsignal said AM message.
 14. A method for mitigating the deleteriouseffects of multipath on a received signal, comprising:a) receiving anamplitude modulated message upon a frequency modulated signal; b)canceling the frequency modulated signal from the received signal andgenerating a raw AM message signal and an instantaneous frequencyestimate signal; c) generating a polynomial with a plurality ofcoefficients for producing an FM to AM conversion interference estimatesignal; d) subtracting the FM to AM conversion interference estimatesignal from the raw AM message signal to produce a clean AM messagesignal; e) measuring and integrating the cross correlation between saidclean AM message signal and each term of the polynomial; and f)adjusting the coefficients of the polynomial, thereby driving the FM toAM interference in the clean AM message signal to zero.
 15. A method forrecovering signals from a transmitted large carrier AM over FM signalincorporating an AM message and an FM message, comprising:a) receivingthe large carrier AM over FM signal; b) generating, from said largecarrier AM over FM signal, a suppressed carrier AM over FM signal; c)generating, from said large carrier AM over FM signal, an FM onlywaveform; d) demodulating the large carrier AM over FM signal, whereinsaid demodulating step comprises combining said suppressed carrier AMover FM signal and said FM only waveform to produce a product signal andgenerating from said product signal said AM message; e) instantaneouslytracking the frequency of said FM only waveform; and f) generating, fromsaid frequency of said FM only waveform, said FM message.
 16. A methodfor mitigating the deleterious effects of multipath on a receivedsignal, comprising:a) receiving an amplitude modulated message upon afrequency modulated signal; b) canceling the frequency modulated signalfrom the received signal and generating a raw AM message signal and aninstantaneous frequency estimate signal; c) generating a runningestimate of an FM to AM conversion interference signal to be expectedfrom the instantaneous frequency estimate signal by comparison of theinstantaneous frequency estimate signal and the raw AM message signalwith data stored in a look-up table; and d) subtracting the FM to AMconversion interference signal from the raw AM message signal to producea clean AM message signal.
 17. A method for recovering signals from atransmitted large carrier AM over FM signal incorporating an AM messageand an FM message, comprising:a) receiving the large carrier AM over FMsignal; and b) isolating an AM waveform from said large carrier AM overFM signal to generate an AM only signal, said isolating stepcomprising:obtaining an FM only signal; and multiplying the largecarrier AM over FM signal with said FM only signal to generate a productsignal; and c) generating from the product signal an envelope of the AMonly signal.